Active filter for reduction of common mode current

ABSTRACT

An active filter for reducing the common mode current in a pulse width modulated drive circuit driving a load, said drive circuit comprising an a-c source, a rectifier connected to said a-c source and producing a rectified output voltage connected to a positive d-c bus and a negative d-c bus, a PWM inverter having input terminals coupled to said positive d-c bus and negative d-c bus and having a controlled a-c output, a load driven by said a-c output of said PWM inverter, a ground wire extending from said load, and a current sensor for measuring the common mode current in said drive circuit in said ground wire, said current sensor producing an output current related to said common mode current, said active filter comprising a first and second MOSFET transistor, each having first and second main electrodes and a control electrode, and an amplifier driving a respective one of the transistors; said first electrode of said first and second transistor coupled to a common node, said second electrodes of said first and second transistors being coupled to said positive d-c bus and said negative d-c respectively; each of said amplifiers having an input coupled to said output of said current sensor and each having an output connected to a respective one of said control electrodes; and a d-c isolating capacitor connecting said common node of said first electrode of said first and second transistors to said ground wire.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a divisional of U.S. patent application Ser. No.09/816,590, filed Mar. 23, 2001, now U.S. Pat. No. 6,636,107, in thename of Brian R. Pelly and entitled ACTIVE FILTER FOR REDUCTION OFCOMMON MODE CURRENT” which application is related to and claims priorityto Provisional Application Serial No. 60/192,976, filed Mar. 28, 2000and No. 60/211,999 filed Jun. 16, 2000.

FIELD OF THE INVENTION

This invention relates to filters for electrical circuits and morespecifically relates to an active filter for reducing or redirecting thecommon mode current in switch mode power supplies and particularly forreducing the common mode current and EMI in a PWM motor drive circuit.

BACKGROUND OF THE INVENTION

High-speed switching devices such as bipolar transistors, MOSFETs andIGBT's enable increased carrier frequency for voltage-source PWMinverters, thus leading to much better operating characteristics.High-speed switching, however, causes the following serious problems,originating from a high rate-of-change in voltage and/or current:

a) ground current escaping to earth through stray capacitors insidemotors and through long cables;

b) conducted and radiated EMI;

c) motor bearing current and shaft voltage; and

d) shortening of insulation life of motors and transformers.

The voltage and/or current change caused by high-speed switchingproduces high-frequency oscillatory common-mode and normal-mode currentswhen the switching device(s) change state because parasitic straycapacitance inevitably exists inside a load, for example, an ac motor,as well as inside the switching converter. Thus, each time an inverterswitching event occurs, the potential of the corresponding inverteroutput terminal moves rapidly with respect to ground, and a pulse ofcommon mode current flows in the d-c link to the inverter, via thecapacitance of the heatsink motor cable and motor windings to ground.The amplitude of this pulse of current for a class B (residential) motordrive is typically several hundred millamps to several amps; and thepulse width is typically 250 to 500 ns. For a class A drive(Industrial), and depending on the size of the motor and length of themotor cable, the pulse current amplitude is typically several ampereswith a pulse width of 250 ns to 500 ns, to 20 amperes or more with apulse width of 1 to 2 micro seconds.

The common mode oscillatory currents may have a frequency spectrum rangefrom the switching frequency of the converter to several tens of MHZ,which creates a magnetic field and will produce radiatedelectro-magnetic interference (EMI) throughout, thus adversely affectingelectronic devices such as radio receivers, medical equipment, etc.

A number of Governmental restrictions apply to the degree of permissibleline current EMI and permissible ground current in certain motorapplications. Thus, in Class B residential (appliances), applications,ground current must be kept below from 1 to 20 mA over a frequency rangefrom 0 to 30 kHz respectively (over a logarithmic curve); and conductedline current EMI must be kept below designated values (less than about60 dBμV) over a frequency range of 150 kHz to 300 MHZ. For motor driveapplications designated as class A Industrial applications, limitationson ground current are less stringent, but line current EMI is stilllimited over the 150 kHz to 30 MHZ range.

Generally, common-mode chokes and EMI filters, based on passiveelements, may not completely solve these problems. Passive filters,consisting of a common mode inductor and “Y” capacitors in the input acline have been used to filter the common mode current in such motordrive circuits. Passive common mode filters may place limits on the PWMfrequency which can be used, are physically large (frequently a majorfraction of the volume of the motor drive structure) and are expensive.Further, they are functionally imperfect in that they exhibit undesiredresonance which runs counter to the desired filtering action. Further,in general purpose industrial drives, the drive circuit and motor areoften connected by cables which are up to 100 meters or more long. Thelonger the cable, the greater the conducted common mode EMI in the motorcable, and the larger the required size of a conventional passive commonmode input filter.

A common-mode transformer with an additional winding shorted by aresistor is known which can damp the oscillatory ground current.Unfortunately, a small amount of aperiodic ground current will stillremain in this circuit.

Active filters for control of the common mode current in a pulse widthmodulated (PWM) controlled motor drive circuit are well known. Suchdevices are typically described in the paper an Active Circuit forCancellation of Common-Mode Voltage Generated by a PWM Inverter, bySatoshi Ogasawara et al., IEES Transactions on Power Electronics, Vol.13, No. 5, (September 1998 and in U.S. Pat. No. 5,831,842 in the namesof Ogasawara et al.

FIG. 1 shows a typical prior art active filter circuit or EMI and noisecanceller for an a-c motor. Thus, in FIG. 1, an a-c source comprising aninput terminal L and a neutral terminal are connected to the a-c inputterminals of a full wave bridge connected rectifier 40. While a singlephase supply is shown, the principles in this and in all Figures to bedescribed can be carried out with a three-phase or multi-phase input.The positive and negative busses of rectifier 40 contain points A and Drespectively and are connected to a three-phase bridge connected PWMcontrolled inverter 41, at inverter terminals B and F. The output a-cterminals of the inverter are connected to a-c motor 42. A filtercapacitor 40 a is also connected across terminals B and F. Motor 42 hasa grounded housing connected to ground wire 43 with ground terminal 43a.

The active filter consists of a pair of transistors Q₁ and Q₂, connectedacross the d-c output lines of rectifier 40 with their emittersconnected at node E. These define amplifiers which are controlled byoutput winding 44 of a differential transformer having input windings 45and 46 connected in the positive and negative output busses of rectifier40. The winding polarities are designated by the conventional dotsymbols. Winding 44 is connected between the control terminals oftransistors Q₁ and Q₂ and the common emitter node E. A d-c isolatingcapacitor 47 is connected to ground line 43 at node C.

The active filter including capacitor 47 defines a path for divertingthe majority of the common mode current which can otherwise flow in thepath L or N, A, B, M (motor 42), 43, 43 a and back to L or N; (or in thereverse path when polarity reverses) or in path L or N, D, F, M, 43, 43a (or in the reverse path when polarity reverses). Thus, most commonmode current can be diverted, for currents from positive terminal A, inthe path B, M, C, E, Q₂, F, B, for “positive current”, and in thepattern B, M, C, E, Q₁, B for “negative” current by the proper controlof transistor Q₁ and Q₂. The path for common mode current flowing intonegative terminal D follows the path F, M, C, E, Q₂, F for “positive”current and F, M, C, E, Q₁, B for “negative” current. The degree ofdiversion depends on the current gain of winding 44 and the current gainof Q₂, for “positive current”, and the current gain of winding 44 andcurrent gain of Q₁, for “negative” current. In order to obtain asufficient degree of diversion of the common mode current, the overallcurrent gain of winding 44 and transistors Q₁ and Q₂ must be high.

The sensing transformer 44, 45, 46 of FIG. 1 has been large andexpensive in order to provide sufficiently high current gain. It wouldbe very desirable to reduce the size and cost of this transformerwithout jeopardizing the operation of the circuit. A further problem isthat because of the high gain required, this closed-loop circuit has atendency to produce unwanted oscillation.

Further, it has been found that the transistors Q₁ and Q₂ may not beable to operate in their linear regions over a large enough range withinthe “headroom” defined by the circuit, thus defeating the activefiltering action. The headroom, or the voltage between the collector andemitter of transistors Q₁ and Q₂ is best understood by considering theapproximate equivalent circuit of FIG. 1, as shown in FIG. 2, in whichthe ground potential at C is the same as that of the neutral line inFIG. 1. Transistors Q₁ and Q₂ are shown as resistors R₁ and R₂respectively with respective parallel connected diodes. The d-c bridge40 is shown as two d-c sources 50 and 51, each producing an outputvoltage of V_(DC)/2 where V_(DC) is the full output voltage between thepositive and negative busses at terminals A and D, and an a-c source 52having a peak a-c voltage of V_(DC)/2.

It can be seen from FIG. 2 that headroom can disappear at differentportions of the cycle of source 52. Thus, consider a first situation inwhich the leakage impedances of transistors Q₁ and Q₂ are the same. Inthis case, the values of resistors R₁ and R₂ in FIG. 2 are about equal.Now, as the ground potential at terminal C swings between (+)V_(DC)/2and (−)V_(DC)/2 with respect to the d-c midpoint at node 53 in FIG. 2,the potential at the emitters of transistors Q₁ and Q₂ also swingsbetween (+)V_(DC)/2 and (−) V_(DC)12, if it is assumed that theimpedance of capacitor 47 is much smaller than R₁ and R₂. Therefore,during the periods when the potential at node E is close or equal to thepotential of the d-c bus (at points B or F), insufficient voltageheadroom exists for the relevant transistors Q₁ or Q₂ to operate aslinear amplifiers, and the active filtering action is lost.

Consider next a condition in which the ground potential at C is the sameas that of the neutral input line N in FIG. 1, and the leakage currentof transistor Q₂ is now much higher than that of transistor Q₁, which,in FIG. 2 would be represented by the condition that R₂ is much lessthan R₁ This then biases the potential at E toward the negative d-c bus(at F). Therefore, the potential at E resides at the negative buspotential for a significant portion of each input cycle. During thisperiod, transistor Q₂ cannot operate as a linear amplifier and theactive filtering action is lost.

Consider next the condition where the ground is at N and the resistanceof transistor Q₁ is much less than that of transistor Q₂; that is, R₁ ismuch less than R₂. This would bias point E toward the positive d-c bus(point B) so that, for a significant portion of each input cycle,transistor Q₁ cannot operate as a linear amplifier.

Lastly, consider a condition in which a small a-c potential (dotted linea-c source 52 a in FIG. 2) exists between the ground wire 43 and neutralN of FIG. 1, as would occur if the grounded neutral of the supplytransformer is electrically remote from the ground connection of themotor drive itself. In that case, the ground potential will swingbetween +(dV+V_(DC)/2) and −(dV+V_(DC)/2) where dV is the peak of thevoltage wave shape of source 52 a in FIG. 2. If the leakagecharacteristics of transistors Q₁ and Q₂ are about equal, the potentialat E will attempt to swing by dV above the positive bus and by (−) dVbelow the negative d-c bus. During these periods, the voltage at E isclamped to the bus voltage by the low impedance reverse characteristicsof transistors Q₁ and Q₂. Thus, no voltage headroom exists for thetransistors to operate during these periods.

It would be very desirable to provide a circuit which providessufficient headroom for transistors Q₁ and Q₂ under the above describedconditions and for clamping the headroom voltages to a prescribedminimum level; or for regulating the average voltage at point E to thed-c midpoint potential.

The active filter of the prior art, as shown in FIG. 1, is alwaysconnected across the full d-c bus voltage. This requires a high enoughvoltage rating for the transistors Q₁ and Q₂ and causes a relativelyhigh power dissipation in the active filter components. It wouldtherefore be desirable to operate the active filter at a lower voltage,if possible, without degrading the performance of the active filter.

SUMMARY OF THE INVENTION

In accordance with a first feature of the invention, an operationalamplifier is used as a buffer/amplifier between the current sensingtransformer and the transistors Q₁ and Q₂. This permits a substantialreduction in the size of the common mode transformer, without affectingthe operation of the circuit.

In accordance with a further feature of the invention, and to ensuresufficient headroom for the transistors Q₁ and Q₂ of FIG. 1, a pair ofseries connected balancing resistors are connected in parallel withrespective ones of transistors Q₁ and Q₂. The novel resistors will havea value which ensures that sufficient voltage headroom is alwaysmaintained for transistor Q₁ and Q₂. This permits a current flow whichis significantly higher than the maximum possible leakage current oftransistors Q₁ and Q₂.

Alternatively, a further novel circuit which has substantially lesspower dissipation while still maintaining sufficient headroom fortransistors Q₁ and Q₂ is provided, using active clamps for the voltageheadroom of the transistors Q₁ and Q₂. In this circuit, theinstantaneous voltage across each of transistors is sensed and comparedto respective references. If the headroom voltage falls below thereference, a feedback error is fed back to the amplifier which drivesthe transistors Q₁ and Q₂ to maintain the required headroom. This novelcircuit substantially reduces the power dissipation needed for voltageheadroom maintenance and reduces the magnitude of ground current at linefrequency through the d-c isolating capacitor 47.

In accordance with a still further feature of the invention, theheadroom voltage control is carried out employing a reference voltagewhich is equal to one-half of the d-c bus voltage. The voltage at theemitter node E is then compared to this reference and the transistors Q₁and Q₂ are controlled by an active regulation scheme which regulates theaverage voltage at E to the d-c midpoint voltage.

In the circuits described above, the amplifiers used need a source ofoperating or control or biasing voltage. A novel floating power supplyis provided which derives its power from the d-c bus voltage and permitsall output control voltages to move dynamically within the positive andnegative bus voltage. This is accomplished by providing respectivecurrent source circuits connected to the positive and negative busses,respectively, and connected to one another through zener referencediodes. The nodes between the current sources and diodes and the nodebetween the diodes form outputs for two control voltages and a commonvoltage reference which all swing dynamically with the bus voltage.

In a still further embodiment of the invention, the operating voltageacross the active filter is sourced from a separate “filter bus” voltagewhich is lower than the full bus voltage. The active filter actsotherwise identically to a circuit driven from the full bus voltage.This novel circuit reduces the power dissipation in the active filtercomponents and lowers the voltage rating of the transistors Q₁ and Q₂.

In accordance with a further important feature of the invention,selected active filter components are integrated into a single siliconchip, defining an active filter IC chip containing the principle activefilter components and having suitable pin-outs for receiving the variousinput and bus connections.

A novel architecture is also provided, enabling the use of a very smalltoroidal current transformer as the common mode sensor. Unlike thearchitecture in FIG. 1, which requires a high gain, this novelarchitecture requires a gain of only unity, and thus avoids the abovementioned problem of unwanted oscillation. The architecture may also beimplemented by MOSFET transistors instead of bipolars for improvedlinearization of the amplifier transfer characteristics wider bandwidthand improved ruggedness.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a circuit diagram of a prior art common mode active filter fora motor drive circuit.

FIG. 2 is an equivalent circuit of the circuit of FIG. 1 to illustratethe issue of headroom voltage.

FIG. 3 is a circuit diagram of one improvement of the invention whichemploys a buffer amplifier between the current sensing transformer andthe transistors.

FIG. 4 shows modification of the circuit of FIG. 3 in which the currentsense primary windings are placed in the a-c circuit.

FIG. 5 is a circuit diagram of a further modification of the circuit ofFIG. 3 in which the primary winding of the current transformer is in theground wire.

FIG. 6 (on the same sheet as FIG. 13) shows an improvement of thecircuit of FIG. 3 which employs balancing resistors across thetransistors to ensure sufficient headroom for their operation.

FIG. 7 shows a further embodiment of a circuit for insuring headroomusing an active clamp circuit.

FIG. 8 is a still further embodiment of a headroom control circuit, inwhich the midpoint d-c bus potential is actively regulated.

FIG. 9 is an equivalent circuit of the circuit of FIG. 8.

FIGS. 10, 11 and 12 show the ground voltage, the voltage acrosstransistors Q₁ and Q₂, the voltage across the filter capacitor and theground current i_(GND) at line frequency for three different headroomconditions.

FIG. 13 (on the same sheet as FIG. 6) shows a novel power supply for theamplifier circuits of the prior figures which derives its power from thed-c bus.

FIG. 14 is a circuit diagram of an improvement of the prior describedcircuits in which the filter bus voltage is less than the main d-c busvoltage.

FIG. 15 is an alternative to the circuit of FIG. 14 in which the reducedfilter bus voltage is derived from a dropping resistor and zener diodecircuit.

FIG. 16 is another circuit for deriving a reduced filter voltage usingseries bus capacitors.

FIG. 17 is a circuit for deriving a reduced filter voltage of one-halfof the full bus voltage and employing a voltage doubler.

FIG. 18 is a circuit similar to that of FIG. 17 in which the filtervoltage is less than one-half of the full d-c bus voltage.

FIG. 19 is a diagram of an integrated circuit chip into which selectedones of the filter components of the preceding figures are integratedinto a single chip of silicon.

FIG. 20 shows the architecture of a circuit having an active filter ICchip in a switching power supply.

FIG. 21 shows a feedback based architecture using an active filter ICchip in a motor drive circuit with the current sensor primary in theground wire.

FIG. 22 shows a modification of the architecture of FIG. 21 employing afeed forward system which permits improved operation.

FIG. 23 shows an architecture like that of FIG. 21, but using adifferential current transformer.

FIG. 24 shows the architecture of FIG. 22 with a differentialtransformer.

FIG. 25 shows the novel active filter of the present invention usingpower MOSFETs instead of bipolar transistors for devices Q₁ and Q₂.

FIG. 26 is an improvement of FIG. 25, adding further control elements.

FIG. 27 shows a modification of the circuit of FIG. 26 for improvedcontrol of the quiescent bias current.

FIG. 28 shows a full circuit diagram of a preferred embodiment of thecommon mode filter of the invention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

FIG. 3 shows an improvement of the circuit of FIG. 1 wherein likenumerals designate similar components as is true throughout thisspecification. FIG. 3 further shows, in schematic fashion, a heat sink49 which receives the IGBTs (or MOSFETs) of the PWM inverter 41. Heatsink 49, like the housing of motor 42, is connected to ground line 43.In FIG. 3, however, a novel improvement is added which permits asubstantial reduction in the size of the transformer 44, 45, 46 withoutaffecting the operation of the active filter function. Thus, in FIG. 3,an operational amplifier 70 is added to the circuit to act as abuffer/amplifier between the secondary of the common mode currentsensing transformer 44, 45, 46 and the transistors Q₁ and Q₂. Thispermits the size and cost of transformers 44, 45, 46 to be substantiallyreduced.

While the primary windings 45 and 46 of the common mode transformer areshown in FIG. 3 at the output of rectifier 40, they may be placed in thea-c input lines as shown in FIG. 4; or, if desired, a single primarywinding 71 can be connected in series with the ground terminal, as shownin FIG. 5. In all cases, the secondary winding 44 is simply coupled toone or more primary windings which carry or otherwise produce a signalrelated to the common mode current.

Regardless of the connection used, the novel circuits of FIGS. 3, 4 and5 will divert the majority of the common mode current flow into theisolator capacitor 47 and transistors Q₁ and Q₂ and away from theexternal ground wire connected to 43 a.

The circuits of FIGS. 1, 3, 4 and 5 will operate as desired providedtransistors Q₁ and Q₂ always have sufficient voltage “headroom” to allowthem to operate as linear amplifiers. However, as pointed out earlierwith reference to FIG. 2, there are situations in which this headroomdisappears during certain portions of the input voltage cycle. Inaccordance with another feature of the invention and as shown in FIG. 6a novel circuit employing balancing resistors will ensure adequateheadroom under these conditions. Thus, in FIG. 6, two resistors 85 and86 are connected across transistors Q₁ and Q₂ respectively. Note thatthe full circuit is not shown in FIG. 6, it is being noted that suchresistors are intended to be connected across the transistors Q₁ and Q₂in FIGS. 1, 3, 4 and 5.

Resistors 85 and 86 must satisfy the following conditions:

1. Their values must be of the same order of magnitude as the reactanceof capacitor 47 to ensure that the peak value of the a-c component ofvoltage at line frequency and developed across transistors Q₁ and Q₂ issubstantially less than one-half the peak d-c bus voltage. For example,capacitor 47 may have a typical value of 0.01 μF and, at line frequency,the resistors 85 and 86 will be 265 kohms each.

2. Resistors 85 and 86 must be sized to carry significantly highercurrent than the maximum possible leakage current of transistors Q₁ andQ₂, in order to “swamp” differences in their leakage currents and ensurean approximately equal voltage balance across the transistors. In aspecific example, in a circuit having a maximum d-c bus voltage of 400volts with transistors Q₁ and Q₂ having a maximum leakage current of 0.5mA each and capacitor 47 having a capacitance of 0.01 μF, the currentthrough resistors 85 and 86 at half the bus voltage, i.e., 200 volts,should be about 2 mA. Thus, resistors 85 and 86 should be about 100kohms. This is the value to be chosen since it is less than the 265 kohmvalue required by criteria 1 above.

The power dissipation in the 100 kohm resistors at the bus voltage of400 volts is about 400 mw. A line frequency component of ground currentof about 0.5 mA will be drawn through resistors 85 and 86 and thecapacitor 47.

As will be later discussed, it would be desirable to integrate thecontrol circuitry for the active filter into an integrated circuit.Thus, the power dissipation should be reduced as much as possible. FIG.7 shows a circuit for the control of headroom which does not requireresistors and which reduces power dissipation to about 20 mw. Thecircuit of FIG. 7 also reduces the line frequency ground current drawnthrough capacitor 47 from 0.5 mA to 0.2 mA.

The principle of operation of the circuit of FIG. 7 is to actively clampthe voltage headroom for transistors Q₁ and Q₂. This is done by sensingthe instantaneous voltage across each of the transistors Q₁ and Q₂ andcomparing it to a reference. If the headroom falls below the referencevalue, an error signal is produced and is fed back to amplifier 70 todrive the base of transistors Q_(1 or Q) ₂ to maintain the desiredheadroom. Thus, transistors Q₁ and Q₂ will carry the common mode currentand will also carry a small added component of current so as to maintaintheir necessary headroom. The power dissipation in each of transistorsQ₁ and Q₂ is only about 20 mw at a bus voltage of 400 V, compared to 400mV for each of resistors 85 and 86 in FIG. 6.

Referring now to FIG. 7, summing circuits 90, 91 and 92 and amplifiers93 and 94 are added as shown. FIG. 7 also shows power supply d-c sources95 and 96 for providing to V_(DD) and V_(SS) inputs respectively tooperational amplifier 70. Each of summing circuits 90 and 91 have oneinput connected to (+)DC Bus and (−)DC Bus respectively, and anotherinput (terminals 97 and 98 respectively) connected to reference voltages(−)ve and (+)ve respectively which are the headroom reference voltagesfor transistors Q₁ and Q₂ respectively. The summed voltages e_(in) ande¹ _(in) respectively of devices of 90 and 91 are applied to operationalamplifiers 93 and 94 respectively which have outputs e_(out) and e¹_(out) which are applied to inputs of summing circuit 92. The linearshapes of the characteristic outputs e_(out) and e¹ _(out) are shown onthe drawing in circled insets. The output of circuit 92 (an error signaloutput) is connected through high frequency filter 99 to the input ofamplifier 70.

Thus, amplifier 70 will amplify the error signal; and amplifiers 93 and94 respectively prevent the headroom voltages for transistors Q₁ and Q₂from falling below their respective reference voltages.

In more detail, +DC BUS to COM_(REF)(−v_(e)) is a negative referencevoltage, which sets the required value of the +DC BUS to COM voltage Thevalue of this reference (−)ve is set to yield the desired headroomclamping level for transistor Q₁, based on the worst case assumptionthat the emitters of transistors Q₁ and Q₂ could instantaneously bepositive with respect to COM by VDD. Thus +DC BUS to COM REF is set torepresent the required minimum headroom for Q₁, plus VDD.

The difference between +DC BUS to COM REF(−ve) and the actual +DC BUS toCOM voltage (+ve) is fed to the input of amplifier 93. The output ofamplifier 93 is zero for positive input, and positive for negative inputvoltage. When the +DC BUS to COM voltage is greater than the absolutevalue of +DC BUS to COM REF, e_(in) is positive, and the output ofamplifier 93 is zero. When +DC BUS to COM attempts to become less thanthe absolute value of +DC BUS to COM REF which is set to represent therequired minimum headroom for Q₁, plus VDD, e_(in) becomes negative, andthe output e_(out) of amplifier 93 becomes positive. This output isinverted at the summing junction 92, then passed through the HF filter99 to the amplifier 70, biasing the output of this amplifier negatively.The result is that transistor Q₂ is biased on just sufficiently that thevoltage headroom across transistor Q₁ is regulated to the set value.

The (−)DC BUS to COM REF is a set positive reference voltage at terminal98. The required value of this reference represents the required minimumheadroom for transistor Q₂, plus the absolute value of VSS.

The difference between (−)DC BUS to COM REF (+ve) and the −DC BUS to COMvoltage (−ve) is fed to the input of amplifier 94. The output ofamplifier 94 is zero for negative input, and negative for positive inputvoltage. When the absolute value of the (−)DC BUS to COM voltage isgreater than (−)DC BUS to COM REF, e¹ _(in) is negative, and the outputof A3 is zero. When the absolute value of the (−)DC BUS to COM voltageattempts to become less than (−)DC BUS to COM REF, e¹ _(in) becomespositive, and the output e¹ _(out) of amplifier 94 becomes negative.This output is inverted at the summing junction 92, then passed throughthe HF filter 99 to the input of amplifier 70, biasing the output ofthis amplifier positively. The result is that transistor Q₁ is biased onjust sufficiently that the voltage headroom across transistor Q₂ isregulated to the set value.

The HF filter 99 removes high frequency components from the errorsignal, caused by the common mode current of the inverter/motor thatflows through CFILT 47. Thus the closed loop regulator corrects againsterrors of potential at E that would otherwise occur at line frequency,but essentially does not attempt to make corrections for fasterinstantaneous deviations due to the switching events of the inverter.

FIG. 8 shows a further alternative to the circuit of FIG. 6 in which thevoltage at node E of the common emitters of transistors Q₁ and Q₂ issensed and the average value of this voltage is regulated to the d-cmidpoint voltage. Thus, in FIG. 8, a reference voltage circuit definingthe d-c mid-voltage consists of resistors 105 and 106 which are lowpower resistors which are of equal value and produce voltage at node 107which is the midpoint of the d-c voltage between the (+)DC Bus and (−)DCBus voltages. This d-c midpoint reference voltage, and the potential atnode E are applied to, and compared by, a summing comparator 108. Thedifference output is then applied to amplifier 109 and its output errorsignal is connected via HF filter 99 as an input to amplifier 70.

In operation, if the voltage at E attempts to rise above the d-cmidpoint reference potential at node 107, amplifier 109 delivers anegative bias signal to the input of amplifier 70. The output ofamplifier 70 also assumes negative bias, turning on transistor Q₂ to theextent necessary to correct the voltage at emitter node E and backtowards the d-c midpoint reference potential.

Conversely, if the voltage at E attempts to fall below the d-c midpointpotential, the output of amplifier 70 assumes a positive bias, turningon transistor Q₁ to the extent necessary to correct the voltage at Eback towards the d-c midpoint potential. The HF filter 99 again removeshigh frequency components from the error signal.

Thus the regulator loop corrects against variations of potential at Ethat would otherwise occur at line frequency, but essentially does notattempt to make corrections for faster instantaneous deviations causedby the switching events of the inverter.

With the scheme of FIG. 8, the average voltage headroom for transistorsQ₁ and Q₂ is maintained at approximately half the bus voltage. Whilethis is more than sufficient headroom, a disadvantage, relative to the“headroom clamping” approach of FIG. 7, is that the line frequencycomponent of voltage across capacitor 47 is increased and thecorresponding line frequency component of ground current that flowsthrough capacitor 47 is relatively high.

FIGS. 9 to 12 illustrate the design trade-offs between headroom voltagefor transistors Q₁ and Q₂ and ground current for the circuits of FIGS.1, 7 and 8 (at 60 Hz).

FIG. 9 is an equivalent circuit of a portion of the circuit of FIG. 8.The a-c voltage source shown in dotted lines between nodes 107 and 43 ais the a-c ground voltage with respect to the DC midpoint referencevoltage. This voltage, as shown in FIG. 10 is one-half of the peak a-cvoltage between a-c lines L and N (assuming a single phase input with Nat ground potential) and this voltage drives ground current throughcapacitor 47. When the circuit is designed to provide generous headroomfor transistors Q₁ and Q₂ as shown in FIG. 10, as by regulating thevoltage at E to the d-c midpoint potential, the ground current i_(GND)at 60 Hz increases as shown in the bottom graph of FIG. 10.

If, however, the voltage at E is allowed to swing freely with respect tothe bus potentials, as shown in FIG. 11, the line frequency voltageacross capacitor 47 is zero and the corresponding line frequency groundcurrent is also zero. The headroom voltage, however, (the center graphof FIG. 11), is insufficient for the desired filtering of the commonmode motor current (as in FIG. 1).

Finally, if, as in FIG. 7, a headroom clamping circuit is employed,allowing the potential at node E to swing freely until the headroomfalls below a set value, the curves appear as in FIG. 12 where headroomfor the transistors is clamped to the set value while the 60 Hz groundcurrent is reduced to a small value during the clamp intervals only.

An isolated floating power supply would normally be provided for theneeded Vdd and Vss supply voltages for the amplifier circuits of FIGS.3, 4, 5, 7 and 8. An obvious way of deriving the required power supplyvoltages would be via a separate isolated winding on the power supplytransformer that serves the control electronics and gate drive circuits.

FIG. 13, however, shows an alternative method for deriving the requiredpower supply voltages. Thus, in FIG. 13, current sources 120 and 121feed a fixed current from the d-c bus, to create the Vdd and Vss powersupply voltages across the regulator zener diodes 122 and 123respectively. The currents of sources 120 and 121 must be equal, andindependent of the voltage at COM. The voltage then moves dynamicallywithin the positive and negative bus voltages, and the headroom iscontrolled as described above.

Note that the current sources 120 and 121 cannot be replaced by simpledropping resistors, because the currents of sources 120 and 121 couldnot then be instantaneously equal, because the two resistors would havedifferent voltages impressed across them as the potential at COMdynamically changes. The difference between such two resistor currentswould be forced through capacitor 47, interfering with the active filterfunction.

The use of the described bus-derived power supply circuit may be limitedby the power dissipation associated with the circuit. This depends uponthe required power supply current, and the bus voltage. For example, ifthe maximum required power supply current is 3 mA, and the bus voltageis 200V, the maximum total dissipation due to the power supply would be600 mW, which is acceptable.

FIG. 14 is a circuit diagram of a novel active filter which has areduced operating voltage. Thus, it has been recognized that it is notnecessary for the active filter to be connected directly across the fulld-c bus voltage. The active filter, complete with the previouslydescribed current source power supply, can be sourced from a separate“filter bus” voltage, that can be lower than the full d-c bus voltage.The headroom control methods already described will still operate in thesame way to maintain the desired headroom for transistors Q₁ and Q₂. Itis necessary only that a low ac impedance path for the common modecurrent is provided from the collectors of both transistors Q₁ and Q₂ toone of the main d-c busses. A low ac impedance path to the other mainbus is then already in place, via the main d-c bus capacitor 40 a.

FIG. 14 shows the arrangement with a separate positive filter bus 130which provides the positive voltage for the active circuit; the negativebus of the active filter being common with the main negative d-c bus.(It is also possible to provide a separate negative filter bus voltage,with the positive bus of the active filter being common with the mainpositive d-c bus). The operational amplifier circuits are schematicallyshown as block 132.

The capacitor 131 provides the required low ac impedance between thepositive filter bus 130 and the main negative bus. Flow paths for thecommon mode current under various operating conditions are the same asthose previously described.

The advantages of using a filter bus voltage that is lower than the maind-c bus voltage are:

(a) a lower power dissipation in the active filter components, includingthe current source power supply.

(b) a lower voltage rating is required for the transistors Q₁ and Q₂,and for the current source transistors of the power supply. A lowervoltage rating for transistors Q₁ and Q₂ is advantageous because thispermits transistors with better Safe Operating Area and better highfrequency performance, giving faster response and improved performanceof the active filter.

The use of a filter bus voltage that is lower than the main d-c busvoltage will reduce the allowed swing of line-frequency voltage acrosstransistors Q₁ and Q₂, because their emitters E must now swing withinnarrower limits. Thus, in order to maintain the required operatingheadroom for transistors Q₁ and Q₂, the line-frequency voltage swingacross filter capacitor 47 may be forced to increase, resulting in anincrease in the line-frequency component of ground current drawn throughcapacitor 47. While this would be the case with single phase input withgrounded neutral; it would not be the case for three phase input withgrounded neutral, or for 1-phase input with midpoint ground. Thus, witha three phase input with neutral ground, the swing of ground voltagewith respect to the d-c midpoint is less than 25% of the main d-c busvoltage. The filter bus voltage would therefore have to be reduced toless than about 25% of the main bus voltage before the line-frequencycomponent of current drawn through capacitor 47 would increase.

FIGS. 15 to 18 show alternate circuits for deriving a lower filter busvoltage from the main d-c bus voltage. In FIG. 15, the filter busvoltage is derived via a dropping resistor 140 and a voltage regulatordiode 141 from the d-c bus. While the resistor 140 dissipates power, itpermits power savings in the active filter itself, and may allowimproved filter performance, by virtue of using lower voltage and widerbandwidth active components.

FIG. 16 employs two series connected d-c bus capacitors 145 and 146,each supporting approximately half the bus voltage. Series connected buscapacitors may be used where the input voltage is 380V or higher. Theprinciple in FIG. 16 is to use the voltage across the lower buscapacitor 146, i.e., approximately half the d-c bus voltage, as thefilter bus voltage. This requires additional active means shown asactive regulator block 147 for keeping the voltages across the two buscapacitors balanced.

FIG. 17 shows an arrangement with a single phase supply L, N input to avoltage doubler circuit. The voltage doubler replaces rectifier 40 ofthe preceding circuits and consists of diodes 160, 161 and capacitors162 and 163. The main d-c bus voltage is approximately twice the peakline voltage. The voltage across each capacitor is half the full d-c busvoltage. The filter bus 130 is connected to the lower bus capacitor 161in accordance with the feature of the invention, and is half the fulld-c bus voltage. No additional means of balancing the voltages acrossthe doubler capacitors is needed, since balancing of these voltages is anatural result of the operation of the voltage doubler circuit.

FIG. 18 is similar to FIG. 17, but adds the resistor 140 and zener diode141 as in FIG. 15 to derive a lower filter bus voltage, which is lessthan one-half of the main d-c bus voltage.

There is next described in FIG. 19 an arrangement for integrating theprior described circuits into an integrated circuit active filter chipproduct.

The integrated circuit of FIG. 19 may have numerous filter componentsintegrated into a single silicon chip, as shown by the dotted lineperiphery 170. The integrated circuit 171 contains transistors Q₁ andQ₂; the voltage divider 105, 106 of FIG. 8; the floating power supply120-123 of FIG. 13; the amplifiers 70 and 109 of FIG. 8 (and a secondamplifier 109 a). The IC chip 170 then has plural pin-outs including thelabeled pins for (+)Filter Bus; (−)Filter Bus; capacitor 47; CT winding44; and capacitor C_(VDD), and C_(VSS). Added pins may be provided fordecoupling capacitors and the like.

FIG. 20 shows how the integrated circuit 170 has a general applicationas an active filter in a switching power supply. Thus, the switchingpower supply may have an input rectifier 40 connected to a-c input linesL and N, and an inverter 41. The output of inverters 41 is connected toa transformer 180 which is then connected to an output rectifier 181.All of parts 40, 41, and 181 are mounted on heat sink 49. The heat sinkis connected to a ground line which contains the primary of a sensingcurrent transformer housing a secondary winding 44, as in FIG. 5.

The active common mode filter integrated circuit 170 is then connectedto the secondary sense winding 44 and controls the common mode currentas previously described.

The use of the current sensor 71, 44 in FIG. 20 is employed rather thanthe differential CT 44, 45, 46 because such a circuit is moreuniversally applicable and advantageous for power supplies and becausethere is no external motor which could have an extraneous ground currentwhich would not be picked up by the current transformer.

In the case of a power supply, all the grounded parts are containedwithin the power supply itself (the output transformer 180 can have aground screen), and, therefore, there may be no stray external groundcurrent.

In the ground connection shown in FIGS. 5 and 20, the CT 71, 44 will besmaller because it has only one primary winding, sized for the groundcurrent, (the ground wire actually must be same size as the “live”wires) versus two primaries, each sized for the full load current. Also,since the ground current is sensed directly, there is no possibility fordifferential errors.

One basic system architecture for the active common mode filter of theinvention is shown in FIG. 21 showing a toroidal current sensingtransformer 200. A basic design objective is that the toroidal currentsensing transformer 200 should be as small as possible. Preferably theprimary should be just a single wire 201 that passes directly throughthe center of the toroidal core, as illustrated in FIG. 21.

The architecture of FIG. 21 is a closed loop feedback based system. Itis a feedback system in that the signal on the primary of the sensor 200is an attenuated signal, so the amplifier consisting of winding 44,buffer 70 and transistors Q₁ and Q₂ must be a high gain amplifier.Consequently, the system is subject to possible oscillation and requiresa moderately large CT. That is, the architecture in FIG. 21 basicallyrequires a high current gain, G, between the output current of theamplifier, i_(o), and the current input, i_(GND), to the primary 201 ofthe current transformer. The reasons for this are: $\begin{matrix}\left. a \right) & {\frac{i_{GND}}{i_{{COM}\quad {MOT}}} = \frac{1}{1 + G}} \\\quad & {{{where}\quad G} = \frac{i_{o}}{i_{GND}}}\end{matrix}$

Therefore, in order to minimize i_(GND), G must have a high value.

b) the propagation delay times of the bipolar transistors, Q₁ and Q₂,result in a lag between the output current i_(o), and the input current,i_(GND), of the CT 200. This lag causes spikes on the ground currentwaveform at the crossover points. This effect becomes more pronounced asthe half-period of the oscillatory current, i_(COM MOT), decreases. Atypical half-period of i_(COM MOT) is 200 to 250 nanoseconds, and thecrossover spikes are significant.

The crossover spikes can be reduced by increasing the gain. The higherthe gain, the more overdrive current is fed to the bases of transistorsQ₁ and Q₂ during the crossover periods, which reduces the lag. The gaincan be increased by increasing the number of primary turns of thecurrent transformer. Unfortunately, this is not desirable because it iscontrary to the basic design objective of minimizing the size of thecurrent transformer and using a primary wire that passes just oncethrough the center of the CT.

In principle, the gain could be increased by increasing the gain of theoperational amplifier circuit 70. In practice, this tends to causeclosed-loop oscillation, which has been found to be difficult tosuppress.

An improved architecture is shown in FIG. 22. The architecture of FIG.22 is a feed forward architecture in which the full forward currenti_(com mot) flows through the primary winding of CT 200 and the fullcurrent is always sensed even as the ground current is being reduced.Consequently, the amplifier needs a gain equal to 1.0. This system, withunity gain, has good stability, and a small CT size. Thus, in contrastto the architecture in FIG. 21, this architecture requires unity currentgain, G, between the current i_(com mot) at the input of the currenttransformer, and the output current, i_(o), of the amplifier.

The ground current, i_(GND), is:i_(GND) = i_(COM  MOT) − i_(o)   = i_(COM  MOT) − G ⋅ i_(COM  MOT)

For perfect cancellation of the ground current, the gain G musttherefore be exactly 1.0.

The fundamental advantages of the architecture of FIG. 22 are:

a) Because the required current gain, G, is only unity, the currenttransformer 200 can be physically much smaller, with just a singleprimary wire passing through the center of the toroidal core.

b) The circuitry has reduced susceptibility to instability, because theoverall current gain, from input of the current transformer 200 to theoutput of the amplifier 170, is only 1.0.

c) A further advantage of the basic architecture of FIG. 22 over that ofFIG. 21 arises where differential primary windings must be used on thecurrent transformer 200, rather than a single ground current sensingprimary.

FIG. 23 shows the same basic feedback system architecture as in FIG. 21for the case of a CT 210 with two differential secondary windings 211and 212 carrying only a small fraction of the common mode current, butcarrying the full normal mode current. Small unbalances between thedifferential windings 211 and 212 will result in imperfect cancellationof the normal mode current, and will yield a relatively large unwantednormal mode signal at the secondary, in addition to the “desired” commonmode signal. The unwanted normal mode signal distorts the desiredfeedback signal, and upsets the operation of the active filter.

FIG. 24 shows the same basic proposed feedforward system architecture asin FIG. 22, and the differential primary windings 211 and 212 of the CT210 carry the full common mode current, as well as the full normal modecurrent. Thus the ratio of unwanted normal mode signal at the secondary44 of the transformer (caused by imperfect cancellation between the twoprimary windings), to the wanted common mode signal, is now much lower,(because the common mode signal is much higher). Any residual normalmode signal thus has a relatively much smaller distorting effect on theoutput of the amplifier.

A potential design problem with the architecture of FIG. 22 is that toobtain good cancellation of the ground current, the input/output currentcharacteristic of the amplifier 170 must be linear, and it must have asclose to unity gain and as small a phase lag as possible. MOSFETs arepotentially better candidates than bipolar transistors for devices Q₁and Q₂, because of their minor propagation delays, as well as theirbetter SOA.

Linearization of the transfer characteristic can be largely achievedthrough the use of MOSFET source followers, in conjunction with astanding d-c bias that offsets the gate threshold voltage. Additional“linearizing” feedback around the amplifier can also be added to correctagainst residual non-linearity and propagation delays. This addedfeedback does not require high gain, therefore it does not inviteclosed-loop instability.

A basic amplifier circuit implementation employing N channel MOSFETtransistor Q₁ and P channel MOSFET transistor Q₂ is shown in FIG. 25. InFIG. 25, components similar to those of the prior Figures have the sameidentifying numeral. A floating power supply 300 is provided to supplybiases V_(DD) and V_(SS). The circuit components are easily integratedinto a chip 170 as shown in dotted line outline. Amplifiers 301 and 302drive N channel MOSFET Q₁ and P-channel MOSFET Q₂ respectively, andprovide respective positive and negative d-c bias voltages that offsetthe gate threshold voltages of the MOSFETs.

Consider amplifier 301, and assume that resistor RFB is open. The outputvoltage of amplifier 301, shown as e_(o), is: $\begin{matrix}{{e\quad o} = {{i_{S}{R_{S}\left( {1 + B} \right)}} + \frac{R_{4}{V_{SS}}}{R_{2}}}} & (1)\end{matrix}$

where i_(S) is the output of winding 44, and Rs, R₁, R₂, R₄, and V_(SS)are shown in FIG. 25, and$B = \left( {1 + \frac{R_{4}}{R_{1}} + \frac{R_{4}}{R_{2}}} \right)$

R₂ is selected so that: ${\frac{R_{4}{V_{SS}}}{R2} = {Vgth}},{where}$

Vgth is the gate threshold voltage of MOSFET Q₁ Thus the output voltageof amplifier 301 has a standing d-c bias that substantially cancels thethreshold voltage of MOSFET Q₁.

The output current, i_(o), of the source follower circuit comprising Q₁,R_(SOURCE), R_(SENSE) is: $\begin{matrix}{i_{o} = {\frac{i_{S}{R_{S}\left( {1 + B} \right)}}{R_{SOURCE} + R_{SENSE}} \times \left( \frac{1 + {{gfs}\left( {R_{SOURCE} + R_{SENSE}} \right)}}{{gfs}\left( {R_{SOURCE} + R_{SENSE}} \right)} \right)}} & (2)\end{matrix}$

If gfs.(R_(SOURCE)+R_(SENSE))>>1, then equation (2) approximates to:$\begin{matrix}{i_{o} = \frac{i_{S}{R_{S}\left( {1 + B} \right)}}{R_{SOURCE} + R_{SENSE}}} & (3)\end{matrix}$

Equation (3) shows that i_(o) is proportional to i_(S). This of courseis based on the assumptions that (a) Vgth is exactly canceled by$\frac{R_{4}{V_{SS}}}{R2}$

and (b) gfs(R_(SOURCE)+R_(SENSE))>>1.

In practice, some degree of non-linearity between i_(o) and i_(S) willoccur. This non-linearity is reduced by the feedback circuit comprisingR_(SENSE) and R_(FB).

R_(SENSE) is chosen so that e_(FB) (=i_(o).R_(SENSE)) is nominally equalto e_(in) (=i_(S).R_(S)). i.e.e_(FB) nominally tracks e_(in). If e_(FB)exactly tracks e_(in), then the voltage across R_(FB) will be zero, nocurrent will flow in R_(FB), and the feedback circuit has no modifyingeffect.

If (i_(o).R_(SENSE)) becomes greater than (i_(S).R_(S)), feedbackcurrent flows through R_(FB), towards the negative input terminal ofamplifier 301; this has the effect of decreasing e_(o), thus reducingthe error.

Conversely, if (i_(o).R_(SENSE)) becomes less than (i_(S).R_(S)),feedback current flows through R_(FB), away from the negative inputterminal of amplifier 301; this has the effect of increasing e_(o), thusreducing the error.

Consider next the CT 200 and designate its numbers of primary andsecondary turns as N_(p) and N_(S) respectively.

Returning to equation (3), and substituting $\frac{N_{p}}{N_{S}}$

i_(COM MOT) for i_(S): $\begin{matrix}{i_{o} = {\left( {\frac{N_{P}}{N_{S}} - \frac{R_{S}\left( {1 + B} \right)}{R_{SOURCE} + R_{SENSE}}} \right)i_{{COM}\quad {MOT}}}} & (4)\end{matrix}$

By appropriate choice of resistor values and N_(S) (secondary turns ofthe CT 200):${\frac{N_{P}}{N_{S}}\frac{R_{S}\left( {1 + B} \right)}{R_{ROURCE} + R_{SENSE}}} = 1.0$

and equation (4) becomes:

i_(o)=i_(COM MOT)

This is the required design condition for i_(GND)=0.

The design principles for the amplifier circuit 302, which drives MOSFETQ2, are similar.

It is desirable that MOSFETs Q₁ and Q₂ are automatically biased, so thata small standing d-c bias current flows through these transistors, fromthe positive d-c bus to the negative d-c bus. This will ensure thatthese transistors are biased just to the point of conduction, therebyminimizing crossover distortion of the output current i_(o).

Some form of closed-loop control of the d-c bias point of MOSFETs Q₁ andQ₂ is thus required, (a) to maintain a small standing bias current,which is sufficient to avoid crossover distortion, but small enough toavoid significant d-c dissipation in MOSFETs Q₁ and Q₂, and (b) tomaintain the potential of the common point of the floating supply 300that powers amplifiers 301 and 302 at approximately the midpointpotential between the positive and negative d-c busses, thus ensuringoperating headroom as well as substantially equal dissipation andvoltage sharing for MOSFETs Q₁ and Q₂.

FIG. 26 shows the added circuits which perform these functions. In FIG.26, amplifiers 311, 312 and 313 are added to the circuit of FIG. 25 andmay be included within the IC chip 170. Thus, referring to FIG. 26,Amplifier 311 senses and amplifies the voltage acrossR_(SOURCE)+R_(SENSE). During “passive” periods, when i_(o) is zero, thevoltage across R_(SENSE) is zero, and the voltage across R_(SOURCE) isdue to standing d-c bias current, i_(BIAS), that flows from the positived-c bus to the negative d-c bus, via MOSFETs Q₁ and Q₂.

During the passive periods, the output voltage of amplifier 311 thusrepresents an amplified inverted version of the voltage acrossR_(SOURCE), which itself represents the standing d-c bias currentthrough MOSFET Q₁. The output voltage of amplifier 311 due to the biascurrent is less than, but close to, the forward threshold voltage ofdiode D1. This diode therefore does not conduct during the passiveperiods.

During the active periods, when output current i_(o) flows throughMOSFET Q₁, the voltage at the output of amplifier 311 quickly tries toexceed the forward threshold voltage diode of D1, but is clamped to thisvoltage. The duty cycle of the active periods is very low relative tothe passive periods, and thus the average output voltage of amplifier311 essentially just represents the standing d-c bias voltage acrossR_(SOURCE), inverted and multiplied by the gain of amplifier 311:${e_{0}(311)} = {{- \frac{R9}{R8}}{i_{BIAS} \cdot R_{SOURCE}}}$

The output voltage of amplifier 312 then becomes: $\begin{matrix}{{e_{0}(312)} = {{{- V_{DD}}\frac{R12}{R11}} - {e_{0{(311)}}\frac{R12}{R10}}}} \\{= {{- \left( {{{e_{0}(311)}{REF}} + e_{0{(312)}}} \right)}\frac{R12}{R10}}} \\{{= {{- \left( {{{e_{0}(311)}{REF}} - {\frac{R9}{R8}{i_{BIAS} \cdot R_{SOURCE}}}} \right)} \cdot \frac{R12}{R10}}}{{{{where}\quad {e_{0}(311)}{REF}} = {V_{DD}\frac{R10}{R11}}},{and}}}\end{matrix}$

e_(o)(311)REF represents the desired fixed reference value for thestanding d-c bias current, i_(BIAS), and;

e_(o(312)), thus represents the amplified error between the desired andactual d-c bias current.

This error voltage is fed to the input of amplifier 301, via R13, suchthat, in addition to its other functions, amplifier 301 regulatesi_(BIAS) to essentially the set reference level. Note that i_(BIAS) mustof necessity also flow in MOSFET Q₂, since the capacitor 47 blocks theflow of any d-c current through R_(SENSE). It remains now to add a meansfor regulating the common point of the floating power supplysubstantially to the midpoint potential between the positive andnegative d-c bus voltages. This function is carried out by amplifier 313which amplifies the difference between the positive and negative busvoltages, sensed by resistors R15 and R16 respectively. The amplifieddifference voltage is fed as an input to amplifier 302, via resistorR14, such that, in addition to its other functions, amplifier 302regulates the common point of the floating power supply substantially tothe midpoint potential between the positive and negative bus voltages.

The novel invention as described above enables the production of a motordrive circuit which can reduce conductive emission noise and meet therequirements of Class A and Class B motor drives. A further advantage ofthe novel invention is that ground leak current is reduced to eliminatefault trip of the circuit due to excessive ground current. The reductionof ground leak current is of great importance for the drive ofcompressor motors using high dielectric constant cooling materials suchas R410A which will have an increased capacitance from motor frame toground.

FIG. 27 shows modifications in the circuit of FIG. 26 in which thefeedback circuit for quiescent bias current is revised; and FIG. 28 is adetailed overall circuit diagram of the common mode filter of theinvention. Referring to FIGS. 27 and 28, the main feedback loop aroundthe amplifiers 301 and 302 is now via R47 and R27 to the sources oftransistors Q₁ and Q₂ respectively. This feedback, directly to thesources of the MOSFETs, essentially forces the voltages e_(o301) ande_(o302) to follow the input voltages e_(in) 301 and e_(in) 302respectively, overcoming the nonlinear transfer characteristics of theMOSFETs Q₁ and Q₂, capacitors C1 and C13 prevent unwanted oscillation ofamplifiers 301 and 302 respectively. Overall linearity between e_(o) ande_(in) for each amplifier is significantly improved, versus thatobtained with the circuit of FIG. 26.

With the above arrangement of feedback resistor connected directly tothe source of the MOSFET, there is no feedback resistor directly aroundthe amplifier itself. Thus the output of amplifier 301 will saturate ife_(in) 301 goes negative, because transistor Q₁ cannot replicatenegative input current, and the output of amplifier 302 would saturateif e_(in) 302 goes positive, because transistor Q₂ cannot replicatepositive input current. Separate “allowed polarity only” inputs arederived via diode D4 and R22 for amplifier 301, and diode D5 and R47 foramplifier 302. e_(in) 301 is positive during the positive period ofi_(com mot), and zero during negative periods. e_(in) 302 is negativeduring negative periods of i_(com mot), and zero during positiveperiods.

Thus the output voltage of amplifier 301 does not saturate wheni_(com mot) is negative, since it receives no input during this period;instead, it stays essentially at the threshold voltage of Q₁. Likewise,the output voltage of amplifier 302 stays essentially at the thresholdvoltage of Q₂ when i_(com mot) is positive. Thus the outputs of each ofthese amplifiers remain at the desired MOSFET gate threshold levelduring their idle periods, ready to drive current in their respectiveMOSFETs at the crossover points of i_(com mot), with minimal crossoverdistortion.

Note that the secondary winding 44 of the current-sensing transformer isessentially a current source. The voltage across R22 is thereforedirectly proportional to i_(com mot), when this is positive. The voltageacross R47 is directly proportional to i_(com mot), when this isnegative. The voltage drops across diodes D4 and D5 do not distort thecurrent waveform, not introduce any significant distortion of thecurrent-dependent signals across R22 and R47.

Amplifiers 311 and 312 in FIG. 26 regulate the quiesent bias current viaresistor R13, amplifier 301 and MOSFET Q₁ The amplifier 313 in FIG. 26regulates the potential at the common point of the source resistors viaresistor R14, amplifier 302 and MOSFET Q₂. These functions are reversedin the circuit of FIG. 28. Amplifiers U2 and U3 (equivalent toamplifiers 311 and 312) regulate the quiescent current via R43 andamplifier U1B (equivalent to 302) and Q₂. Amplifier U4 (equivalent to313) regulates the potential at the common point of the source resistorsvia resistor R7, amplifier U1A (equivalent to 301) and Q₁. The swappingof these functions between Q₁ and Q₂ does not change the basic principlebut the arrangement of FIG. 28 has been found to be better in practice.

With reference to FIG. 26 the clamping diode D1 across the amplifier 311is one way of mitigating unwanted output voltage of amplifier 311 whenoutput current flows through Q₁ to ground via 47. Ideally, however, theoutput of amplifier 311 should always represent just the quiescent biascurrent, without any superimposed component of the output current, andshould not change during the pulses of output current. In practice, thismethod of FIG. 26 of “clamping out” the output current component fromthe quiescent current feedback signal at the output of amplifier 311generally does not allow sufficiently accurate regulation of thequiescent bias current, because the quiescent bias current feedbacksignal is corrupted to some degree during each pulse of output current.

In the circuits of FIGS. 27 and 28, components R50, R51, R52, D10 andd11 are added to solve this problem. Thus, voltage proportional to theoutput current carried by transistor Q₂ is developed across R50. Thisvoltage is applied as a second input to amplifier 311/U2, via R52, sothat it exactly cancels the voltage developed across R_(SOURCE (Q2))(R30 etc), caused by the output current component, which is applied tothe input of amplifier 311 (U2) via R12. The net input to amplifier 311(U2), via the combination of R12 and R52, is thus due only to thequiesent bias current that flows through R_(SOURCE (Q2)) and Q₂. It doesnot flow at the output into R50 or R51. The output of 311 (U2) now staysconstant at a level that represents just the quiescent bias current,whether or not output current is flowing. With this arrangement, theclamping diode D1 in FIG. 26 is no longer needed.

Although the present invention has been described in relation toparticular embodiments thereof, many other variations and modificationsand other uses will become apparent to those skilled in the art. It ispreferred, therefore, that the present invention be limited not by thespecific disclosure herein.

What is claimed is:
 1. An active filter for reducing the a common modecurrent in a pulse width modulated drive circuit driving a load; saiddrive circuit comprising an a-c source, a rectifier connected to saida-c source and producing a rectified output voltage connected to apositive d-c bus and a negative d-c bus, a PWM inverter having inputterminals coupled to said positive d-c bus and negative d-c bus andhaving a controlled a-c output, a load driven by said a-c output of saidPWM inverter, a ground wire extending from said load, and a currentsensor for measuring the common mode current in said drive circuit insaid ground wire, said current sensor producing an output currentrelated to said common mode current; said active filter comprising afirst and second MOSFET transistor, each having first and second mainelectrodes and a control electrode, and an amplifier driving arespective one of the transistors; said first electrode of said firstand second transistor coupled to a common node, said second electrodesof said first and second transistors being coupled to said positive d-cbus and said negative d-c respectively; each of said amplifiers havingan input coupled to said output of said current sensor and each havingan output connected to a respective one of said control electrodes; anda d-c isolating capacitor connecting said common node of said firstelectrode of said first and second transistors to said ground wire. 2.The active filter of claim 1, wherein the amplifiers provide respectivebias voltages that offset gate threshold voltages of each of thetransistors.
 3. The active filter of claim 2, wherein one transistor isan N channel MOSFET and the other is a P channel MOSFET and therespective bias voltages comprises positive and negative d-c voltages.4. The active filter of claim 1, further comprising a feedback circuitcoupling the common mode node to an a respective input of saidamplifiers.
 5. The active filter of claim 4, wherein the feedbackcircuit comprises a feedback resistor coupling the common node torespective inputs of said amplifiers.
 6. The active filter of claim 5,wherein the feedback circuit further comprises a further resistorcoupling said common node to a common connection, said d-c isolatingcapacitor being coupled between said common connection and said groundwire.
 7. The active filter of claim 6, wherein the common connectioncomprises a common point of a floating power supply that providespositive and negative voltages to respective one of said amplifiers. 8.The active filter of claim 7, wherein the common point of the floatingpower supply is approximately at a midpoint potential between thepositive and negative d-c busses.
 9. The active filter of claim 8,further comprising an amplifier stage coupled to at least one of saidtransistors for providing a feedback voltage to at least one of saidamplifiers to regulate a bias current flowing through at least one ofsaid transistors to a reference level.
 10. The active filter of claim 8,further comprising an amplifier stage having an input coupled to saidpositive and negative d-c busses and an output connected to an input ofat least one of said amplifiers, thereby to regulate the common point ofthe floating power supply substantially to the midpoint potentialbetween the positive and negative d-c bus voltages.
 11. The activefilter of claim 10, wherein the amplifier stage is connected to thepositive and negative d-c busses by a resistive voltage dividerconnected between the busses.
 12. The active filter of claim 4, whereinthe sources of said first and second transistors are coupled togethervia respective feedback resistors to the common node.
 13. The activefilter of claim 12, wherein the feedback resistors are coupled directlyto the common node.
 14. The active filter of claim 12, wherein thefeedback resistors are coupled to the common node via the respectiveresistors, and the feedback resistors are coupled respectively to thesources of the first and second transistors, thereby forcing a voltageacross the respective resistors to follow respective input voltages tothe amplifiers driving respective ones of the transistors.
 15. Theactive filter of claim 14, further comprising oppositely polarizeddiodes coupling the current sensor output and respective inputs of theamplifiers.
 16. The active filter of claim 14, further comprisingoppositely polarized diodes coupling the common node to said isolatingcapacitor.
 17. The active filter of claim 9, wherein the amplifier stageincludes an amplifier having a clamping diode coupling an input andoutput of the amplifier, thereby clamping the output voltage of theamplifier to the input voltage of the amplifier when current flowsthrough at least one of said transistors.
 18. The active filter of claim16, wherein the oppositely polarized diodes coupling the common node tothe isolating capacitor are coupled to the common node by respectivecoupling resistors.
 19. The active filter of claim 18, furthercomprising a first resistor coupling the source of one of saidtransistors to the input of said amplifier stage, and further comprisinga second resistor coupling a voltage developed across one of saidcoupling resistors to said input of said amplifier stage, therebycanceling a voltage caused by a current flowing in one of saidtransistors such that the output of the amplifier stage stays constantat a level representing a quiescent bias current through at least one ofsaid transistors.